Device and method of commutation control for an isolated boost converter

ABSTRACT

A device and method of commutation control for an isolated boost converter provides a unique commutation logic to limit voltage spikes by utilizing switches on the secondary side to minimize a mismatch between current in the inductor and current in the leakage inductance of the transformer when commutation takes places. To minimize this mismatch, the current in the leakage inductance is preset at a certain level that approaches the current in the inductor prior to the commutation, thus significantly reducing the power rating for a clamp circuit and enabling use of a simple passive clamp circuit. In addition, through unique timing of the turn-on of the secondary switches, soft switching conditions are created that eliminate turn-on losses and the reverse recovery problems of free-wheeling diodes.

CROSS-REFERENCE TO RELATED APPLICATION Cross Reference to RelatedApplications

[0001] This application claims the benefit under 35 U.S.C. §119(e) ofthe U.S. Provisional Patent Application No. 60/319,070 filed Jan. 16,2002, entitled Device And Method Of Commutation Control For An IsolatedBoost Converter, such application hereby incorporated by reference inits entirety.

BACKGROUND OF THE INVENTION

[0002] 1. Field of the Invention

[0003] The present invention relates generally to the field of directcurrent-to-direct current (DC/DC) converters, and more particularly tocommutation control schemes for an isolated boost converter.

[0004] 2. Description of the Related Art

[0005] Isolated DC/DC converters for converting a low voltage directcurrent (dc) power source, such as a 12 volt battery, to a high voltageDC power source, such as a 300V traction battery are known in the art.An example of such a converter in the form of an isolated boost DC/DCconverter is illustrated in FIG. 1. In such a converter, an inductorL_(f) is used as the current source at the low voltage side V_(b) toreduce the RMS (root-mean-square) current rating of low voltagetransistors S₁, S₂, S₃, and S₄. The low voltage transistors S₁, S₂, S₃,and S₄ operate as an inverter to convert DC current (voltage) to a highfrequency alternating current (ac) (voltage). An isolation transformer Tsteps up the voltage to a higher level by the turns ratio, whileproviding galvanic isolation for safety regulations. Diodes D₅, D₆, D₇,and D₈ operate as a rectifier to convert the high frequency AC current(voltage) to the desired high DC voltage.

[0006] Referring to FIG. 1, on the input side, the function is basicallyto chop the low voltage from the V_(b) energy source, such as the 12volt battery voltage, into the AC voltage. For this particulararrangement, the inductor L_(f) is provided to limit the inrush current.Thus, the inductor is provided to regulate the input current to limitthe current from the battery. This has a number of advantages, a primaryone of which is that smaller devices such as the switches, S₁, S₂, S₃,and S₄, with a lower current rating, can be used. Therefore, anadvantage of adding the inductor is to limit the current, so thatsmaller devices can be utilized. A problem in using the inductor is thatthe converter also includes the transformer T, which posses non-zeroleakage inductance L_(lk). As more fully explained below, this non-zeroleakage inductance creates a problem whenever the switch states arechanged.

[0007] The way in which the primary side of the converter generates a 12volt plus or minus square wave is that first, switches S₁ and S₂ areturned on. This connects the A terminal of the transformer to thepositive or P battery terminal and the B terminal of the transformer tothe negative or N battery terminal. Thus, if the voltage V_(ab) acrossterminals A and B is plotted as a function of time, the V_(ab) will beplus 12 volts. Thereafter, switches S₁ and S₂ are turned off, andswitches S₃ and S₄ are turned on instead. Basically, the polarity ofV_(ab) is thereby reversed, and the V_(ab) becomes negative 12 volts.Continuing to alternate the switches in this way produces a squarewaveform, and the DC voltage is changed to AC voltage. Thus, switchesS₁, S₂, S₃, and S₄ invert the DC voltage to AC voltage and are referredto collectively as the inverter.

[0008] Referring once more to FIG. 1, when switches S₁ and S₂ are on,the current is drawn from the battery V_(b) and goes through theinductance L_(f) into the leakage inductance L_(lk) of the transformerand then flows out of terminal B and is returned through switch S₂ tothe battery V_(b). When the polarity is changed by turning off switchesS₁ and S₂ and turning on switches S₃ and S₄, the current likewise goesthrough the inductance L_(f). However, the current then flows throughswitch S₄ to terminal B first and thereafter through the leakageinductance L_(lk) of the transformer and out from the leakage inductanceL_(lk) to terminal A and is then returned through switch S₃ to thenegative bus N. Thus, it can be seen that the current inside the leakageinductance L_(lk) has a reversed polarity. This process is referred toas commutation.

[0009] Whenever there is a change of current in an inductor, there is avoltage across the inductor referred to as Ldl/dt, i.e., the inductanceL times the rate of current change dl/dt. The change of the polarity ofthe current in the leakage inductance occurs as the switches are turnedon and off at a level of micro-seconds. Therefore, if the input currentis, for example, 150 amperes, it becomes negative 150 amperes in one ortwo micro-seconds. Thus, the dl/dt is very high, and if the inductanceleakage is not at zero, the voltage can be sizeable. For example, theleakage inductance associated with the power stage of a DC/DC converteris typically in the range of ten microhenries. When the current isreversed 300 amperes from plus 150 amperes to negative 150 amperes inhalf a micro-second, if the leakage inductance is, for example, four orfive microhenries, the voltage is in the range of about 2000 volts. Theswitches S₁, S₂, S₃, and S₄, which are power transistors, have a voltagelimit, and it can be very difficult for these transistors to withstandsuch a high voltage spike.

[0010] This huge voltage spike can damage the power transistor S₁, S₂,S₃, and S₄. A typical approach to dealing with this problem is a passiveclamp circuit, as shown in FIG. 1, which is basically a capacitor with adiode. When the voltage is beyond the capacitor voltage, the diodeconducts and diverts the energy to the capacitor to clamp out thevoltage spike. In that way, the switches S₁, S₂, S₃, and S₄ can beprotected from an over-voltage situation. However, each time the voltageis clamped, energy that is transferred from the leakage inductanceL_(lk) is stored in the clamp capacitor. That energy will charge up thecapacitor unless a way is provided to deplete the capacitor or toconsume the energy each time the transistors are switched. That isusually done by a resistor in parallel with the capacitor to bleed outthe capacitor voltage. Each time the polarity of the current in theleakage inductance L_(lk) is switched, the energy stored in the leakageinductor is lost and therefore wasted if a passive clamp circuit isused.

BRIEF SUMMARY OF THE INVENTION

[0011] It is a feature and advantage of the present invention to providea device and method of commutation control for an isolated boostconverter that minimizes energy that is wasted in a clamp circuit bycreating a momentary short circuit on the secondary winding of thetransformer, thereby improving efficiency and minimizing the size of theclamp circuit.

[0012] It is a further feature and advantage of the present invention toprovide a device and method of commutation control for an isolated boostconverter that creates proper switching conditions to eliminateoscillation due to parasitic parameters by appropriate timing of theturn-on of secondary switches.

[0013] To achieve the stated and other features, advantages and objects,an embodiment of the present invention provides a soft switchingcommutation control scheme for isolated boost converters in whichcurrent in the leakage inductance of the transformer is preset tosignificantly reduce the mismatch current with the inductor duringcommutation, thereby significantly reducing the power rating for thevoltage clamp circuit for the low voltage inverter. Thus, a simplepassive clamp circuit becomes feasible. A unique control logicimplements a partial resonant and then freewheeling operation from thehigh voltage side to achieve leakage current presetting. This controlscheme makes use of parasitic parameters, such as transformer leakageinductance and transistor output junction capacitors, and the operationis insensitive to these parasitic parameters. Natural soft switching isachieved for the transistors at the high voltage side, which leads tolower switching loss and lower electromagnetic interference (EMI).

[0014] An embodiment of the present invention utilizes, for example, aninductor capable of storing energy from a direct current power source,such as a DC battery, an inverter circuit coupled to the inductorcapable of converting direct current to alternating current, atransformer coupled to the inverter and having a primary and a secondaryand capable of stepping up voltage of the alternating current, and arectifier circuit coupled to the transformer that is capable ofconverting the stepped up alternating current to direct current fordelivery to a load. In addition, a programmable controller ispre-programmed with a commutation logic whereby, in a charging inductorswitching mode of the inverter circuit that is timed to occur betweenalternating positive and negative current switching modes of theinverter circuit, energy from the direct current power source is storedin the inductor, and the leakage inductance current of the transformeris preset to a value close to the inductor current by short circuitingthe transformer secondary during the charging inductor switching mode ofthe inverter circuit.

[0015] The inverter circuit includes, for example, first, second, thirdand fourth switches (S₁, S₂, S₃, and S₄), and the programmablecontroller is programmed with commutation logic whereby, in the charginginductor switching mode of the inverter circuit, the primary is shortcircuited with the first, second, third and fourth switches (S₁, S₂, S₃,and S₄) turned on. In the positive current switching mode, the first andsecond switches (S₁ and S₂) are turned on and the third and fourthswitches (S₃ and S₄) are turned off, energy is transferred from theinductor to the secondary, arid current in the transformer is positive.In the negative current switching mode, the third and fourth switches(S₃ and S₄) are turned on and the first and second switches (S₁ and S₂)are turned off, energy is transferred from the inductor to thesecondary, and current in the transformer is negative.

[0016] The rectifier circuit includes, for example, fifth, sixth,seventh, and eighth switches (S₅, D₆, S₇, and D₈), and the programmablecontroller is also programmed with commutation logic whereby the leakageinductance current of the transformer is preset to the value close tothe input inductor current and the commutation of the leakage inductancecurrent is accelerated. This is achieved by turning on either the fifthor seventh switches (S₅ and S₇) to short circuit the secondary duringthe charging inductor switching mode of the inverter circuit. The sixthand eighth switches (D₆ and D₈), for example, are diodes. D₈ isautomatically on when S₅ is turned on, while D₆ is automatically on whenS₇ is turned on. Either the fifth (S₅ with D₈ automatically on) orseventh switch (S₇ with D₆ automatically on) is turned on during thepositive and negative current switching modes to create a momentaryshort circuit on the secondary of the transformer to preset the leakageinductance current in the transformer to a predetermined value thatapproaches a current value in the inductor.

[0017] In the soft switching aspect, the controller is programmed withcommutation logic whereby the fifth (S₅) and seventh (S₇) switches areeach turned on according to a predetermined time sequence to create zerovoltage switching conditions. To accomplish this, the fifth (S₅) andseventh (S₇) switches are each turned on into a conducting diode. Thezero voltage switching eliminates turn on loss and eliminates thereverse recovery problems for freewheeling diodes.

[0018] Additional novel features, advantages and objects of theinvention will be set forth in part in the description which follows,and in part will become more apparent to those skilled in the art uponexamination of the following, or may be learned by practice of theinvention.

BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS

[0019]FIG. 1 is a schematic diagram that illustrates an example of atypical isolated boost DC/DC converter for converting a low voltagedirect current (dc) power source to a high voltage DC power source.

[0020]FIG. 2 is a plot which shows examples of waveforms for theoperation of the DC/DC converter of FIG. 1.

[0021]FIG. 3 is a schematic diagram that illustrates an example of theisolated boost topology for an embodiment of the present invention.

[0022]FIG. 4 is a plot which shows examples of waveforms for theoperation of the DC/DC converter control scheme of FIG. 2.

[0023]FIG. 5 is schematic diagram that illustrates an example of theDC/DC converter control scheme of FIG. 2 in which diodes D₆ and D₈ arereplaced with transistors to achieve bi-directional operation.

DETAILED DESCRIPTION OF THE INVENTION

[0024] In the following description, certain specific details are setforth in order to provide a through understanding of various embodimentsof the invention. However, one skilled in the art will understand thatthe invention may be practiced without these details. In otherinstances, well-known structures associated with electrical circuits andcircuit elements have not been shown or described in detail to avoidunnecessarily obscuring descriptions of the embodiments of theinvention.

[0025] Unless the context requires otherwise, throughout thespecification and claims which follow, the word “comprise” andvariations thereof, such as, “comprises” and “comprising” are to beconstrued in an open, inclusive sense, that is as “including, but notlimited to.”

[0026] Reference is now made in detail to an embodiment of the presentinvention, an example of which is illustrated in the accompanyingdrawings, in which like numerals designate like components. The presentinvention makes use, for example, of isolated buck DC/DC convertertopology to run a dual mode-boost mode operation with a novel softswitching commutation control scheme. FIG. 3 is a schematic diagram thatillustrates an example of the isolated boost topology enabled by anembodiment of the present invention. Referring to FIG. 3, the isolatedboost DC/DC converter topology of the present invention includes, forexample, the input source V_(b) 10, the input inductor L_(f) 12, thepositive and negative bus P 14 and N 16, the low voltage transistors S₁18, S₂ 20, S₃ 22, and S₄ 24, and the isolation transformer 26. As usedherein, on the secondary side in FIGS. 1, 3, and 5, unless otherwiseprovided, the letter “S ” indicates an active switch, the letter “D”indicates a diode, and the numerals 5 through 8 indicate a position.

[0027] An embodiment of the present invention includes, for example, twoaspects. One aspect involves creating a momentary short circuit on thesecondary winding 28 of the transformer 26 to minimize the energy goinginto the clamp circuit 30, thereby improving efficiency and minimizingthe size of the clamp circuit 30. Another aspect involves properlytiming the turn-on of the secondary switches S₅ 32 and S₇ 36 to createproper zero voltage switching conditions that eliminate oscillation dueto parasitic parameters and turn on losses. According to an embodimentof the present invention, a unique commutation logic is utilized tolimit the voltage spike by utilizing high voltage switches S₅ 32, D₆ 34,S₇ 36, and D₈ 38. This is accomplished, for example, by minimizing themismatch between the current in the inductor L_(f) 12 and the current inthe leakage inductance L_(lk) 40 when commutation takes place. In orderto achieve this minimization of mismatch, the current in the leakageinductance L_(lk) 40 is preset at a certain level that is close to thecurrent in the inductor L_(f) 12 prior to the commutation. Consequently,the power rating for the clamp circuit is significantly reduced and asimple passive clamp circuit 30, as shown in FIG. 3, can be employed.

[0028]FIG. 4 shows waveforms for operation of the DC/DC convertercontrol scheme of FIG. 3. The low voltage inverter 42 consisting ofswitches S₁ 18, S₂ 20, S₃ 22, and S₄ 24, operates basically in threemodes, including a 4ON mode 44 (also referred to as the charging mode),a S₁ and S₂ ON mode 46 (also referred to as the positive mode), and a S₃and S₄ ON mode 48 (also referred to as the negative mode). For the 4ONmode 44, all four of the transistors S₁ 18, S₂ 20, S₃ 22, and S₄ 24, areturned on. In the 4ON mode 44, the energy is transferred from the lowvoltage power source V_(b) 10, such as a battery, to the inductor L_(f)12 (charging the inductor). For the S₁ and S₂ ON mode 46, the energystored in the inductor L_(f) 12 is transferred to the high voltage side50 via the transformer 26, and current in the transformer 26 is positive(I_(p) 52). For the S₃ and S₄ ON mode 48, the energy stored in theinductor L_(f) 12 is transferred to the high voltage side 50 via thetransformer 26, and current in the transformer 26 is negative (I_(p)52). The commutation takes place when the operation changes among thethree basic modes, 4ON 44, S₁ and S₂ ON 46, and S₃ and S₄ ON 48.

[0029] In a conventional converter as shown in FIG. 1, the outputrectifier bridge comprises four diodes D₅, D₆, D₇, and D₈. Referring toFIG. 2 for current waveforms of the conventional converter duringcommutation, during the 4ON mode (prior to t₀ and from t₂ to t₃) theprimary of the transformer is disconnected from the input inductor L_(f)and short circuited. As a result, the transformer leakage inductancecurrent I_(l) settles to zero after some initial oscillations. On thesecondary side, all the diodes cease to conduct after the leakageinductance current reaches zero. When the low voltage inverter on theprimary side switches from 4ON mode to S₁ and S₂ ON mode, the leakageinductance is connected to the input inductance. The leakage inductancecurrent I_(p) will ramp up from zero to the inductor current IL which isroughly the same as the load current. This transition is completed inthe commutation period t₀ to t₁. As soon as current in the leakageinductance starts to flow, the diodes D₅ and D₆ on the secondary of thetransformer start to conduct. In this case, the output voltage on thesecondary of the transformer, U_(cd), equals the output voltage V_(o).Similarly, when the low voltage inverter switches from 4ON mode to S₃and S₄ ON mode, at t₃, the leakage inductance current will ramp downfrom zero to the negative of the inductor current I_(L). In this case,D₇ and D₈ conduct and U_(cd)=−V_(o).

[0030] The difference between the leakage inductance current I_(p) andthe input inductor current ¹L during the commutation interval is thecurrent flowing into the clamping circuit I_(C). Referring again to FIG.2, the shaded area beneath the clamping current I_(C) represents thetotal charge transferred into the clamp capacitor C_(c) each timecommutation occurs. The total charge multiplied by the clampingcapacitor voltage V_(c) represents the energy being transferred to theclamping capacitor each time commutation occurs. As discussedpreviously, the energy transferred to the clamping circuit is eventuallyconsumed by the resistor R_(c). A feature and advantage of an embodimentof the present invention is to minimize the commutation energytransferred to the clamping circuit.

[0031] One major drawback of the conventional boost converter in FIG. 1is that it results in excessive commutation energy loss in the clampingcircuit. This is due to two reasons. First, the leakage inductancecurrent I_(p) is always reset to zero prior to the commutation period.As a result, the peak value of the clamping current is always I_(L),which is roughly equal to the load current. For a 3 kW boost converter,the peak I_(C) can be as high as 350A. This excessively high peak I_(C)yields excessive commutation loss in the clamping circuit.

[0032] The second reason why commutation loss is high in theconventional boost converter is due to the relatively long commutationinterval, i.e. t₀ to t₁ or t₃ to t₄ in FIG. 2. Referring to theconventional circuit in FIG. 1, at to, S₁ and S₂ on the primary side ofthe transformer is turned on. The primary side voltage U_(ab) is quicklybuilt up to the clamping voltage V_(c). Usually, V_(c) is selected to beclose to the input voltage V_(b) (for example, V_(c)=2V_(b)) so as tominimize the voltage rating of the switches S₁, S₂, S₃, and S₄. On thesecondary side, D₅ and D₆ are on, imposing +V_(o) to the secondary ofthe transformer. When the secondary voltage is reflected by the turnsratio, n_(T), to the primary, the reflected voltage V_(o)/n_(T) isapproximately equal to V_(b). The voltage across the leakage inductance,L_(lk), is the difference between the clamp voltage and the reflectedsecondary voltage, i.e. V_(c)-V_(o)/n_(T), which is a relatively smallquantity. Since the voltage across the leakage inductance L_(lk) isproportional to the rate of change of the current in L_(lk), it followsthat a small voltage across L_(lk) leads to the slow change of thecurrent in L_(lk), i.e. it would take a relatively long time for thecommutation to complete. The long commutation time means a large valuefor the shaded area in FIG. 2, which means more commutation energy loss.

[0033] A feature of the present invention involves reducing thecommutation energy loss by overcoming the aforementioned difficulties inthe conventional boost converter. In this aspect, two diodes on thesecondary side of the transformer, for example, D₅ and D₇ shown in FIG.1, are replaced by two active switches S₅ 32 and S₇ 36 as shown in FIG.3. Referring to FIG. 4 for the operation of the boost converter for anembodiment of the present invention, from t₃ 84 to t₄ 54, the primaryside inverter 42 is in S₁, S₂ ON mode 46. On the secondary side 50, theswitch S₅ 32 and diode D₆ 34 are on, the energy is transferring fromV_(b) 10 to V_(o) 68. S₅ 32 is turned on sometime between t₃ 84 and t₄54. However, because of the direction of the secondary current, S₅ 32 isnot carrying the current. At t₄ 54, S₃ 22 and S₄ 24 are turned on. Theprimary side inverter 42 enters 4ON mode 44. During this mode, theprimary of the transformer 26 is shorted and disconnected from the inputinductance 56. Consequently, the current I_(p) 52 in the leakageinductance decays rapidly from I_(L) 72 towards zero. A resonant circuitexists between the leakage inductance L_(lk) 40 and the parasiticcapacitance associated with the semiconductor switches S₅ 32, S₇ 36, D₆34, and D₈ 38, which push the leakage inductance current I_(p) 52 tonegative value. Once the current I_(p) 52 becomes negative, it flowsthrough D₈ 38 and S₅ 32 which has already been turned on. S₅ 32 and D₈38 effectively short circuit the secondary of the transformer 26,providing zero voltage across the leakage inductance L_(lk) 40. As aresult, the leakage inductance current 52 is “held” at—I_(pk) 58, whichis determined by the output voltage V_(o) 68, the leakage inductanceL_(lk) 40 of the transformer 26, and the parasitic capacitors of theswitches, during the entire 4ON mode 44.

[0034] Referring further to FIG. 4, at t₆ 62, S₁ 18 and S₂ 20 are turnedoff, the primary side inverter 42 enters the S₃, S₄ ON mode 48. Theleakage inductance current I_(p) 52 is expected to build up to I_(L) 72.Since the leakage inductance current I_(p) 52 has been held at I_(pk) 58in the 4ON mode 44 prior to the commutation, the mismatch between thecurrent in the input inductor 56 and the current I_(p) 52 in the leakageinductance is minimized. Therefore, the peak value of the currentflowing into the clamping circuit, I_(Cc) 98, is minimized.

[0035] Moreover, in an aspect of the present invention, the commutationinterval is shortened. Referring again to FIG. 4, S₅ 32 remains on untilt₈ 86. This means that the transformer secondary remains shorted duringthe entire commutation interval from t₆ 62 to t₈ 86. As a result, thereflected voltage to the primary is zero. The voltage across the leakageinductance L_(lk) 40 during the commutation interval is the clampingvoltage V_(c). In comparison to the conventional DC/DC case, where thevoltage across the leakage inductance L_(lk) 40 during the commutationinterval is V_(c)-V_(o)/n_(T), the present invention provides highervoltage across the leakage inductance L_(lk) 40, thereby acceleratingthe commutation process. This shortened commutation interval, combinedwith lower peak value of I_(Cc) 98, results in significant reduction inthe commutation energy loss.

[0036] In a symmetrical manner, one can analyze the operation of thecircuit during the time intervals of t_(g) 88 to t₁₀ and t₁ 82 to t₃ 84with proper switching pattern for S₇ 36.

[0037] In addition to the advantage of reduced commutation loss, anembodiment of the present invention provides an additional feature ofsoft switching. Referring once more to FIG. 4, S₅ 32 is turned on at orbefore t₄ 54 when its anti-parallel diode is actually conducting thecurrent. A zero voltage turn-on condition is created for S₅ 32.Consequently, there is essentially no turn-on loss for S₅ 32. Also, whenS₅ 32 is turned on, its complementary diode, the anti-parallel diode forS₇ 36, is not conducting current. Therefore, reverse recovery of thediode in S₇ is not an issue here. Hence, a slow diode such as the bodydiode of a MOSFET can be used instead of expensive power diodes withfast recovery characteristics.

[0038] An embodiment of the present invention includes, for example, atleast two important aspects. One such aspect is to create a momentaryshort circuit on the secondary winding 28 to minimize the energy goinginto the clamp circuit 30, thereby improving efficiency and minimizingthe size of the clamp circuit 30. Another aspect is properly timing theturn-on of the secondary switches S₅ 32 and S₇ 36 to create properswitching conditions that eliminate turn on losses in S₅ 32 and S₇ 36,and eliminate the need for special fast recovery diodes.

[0039] In an embodiment of the present invention, active switches S₅ 32and S₇ 36, as shown in FIG. 2, are used to replace diodes D₅ and D₇ ofthe conventional rectifier as shown in FIG. 1. Further, four differentarrangements of the power switching devices on the secondary of thetransformer can produce identical results, namely, active switches S₅and S₇ and diodes D₆ and D₈; active switches S₅ and S₈ and diodes D₆ andD₇; active switches S₆ and S₇ and diodes D₅ and D₈; and active switchesS₆ and S₈ with diodes D₅ and D₇, according to an embodiment of thepresent invention.

[0040] Referring once more to FIG. 3, controller unit 300 controlsswitches S₁-S₇ in the manner described herein (e.g., as described inrelation to FIG. 4), according to an embodiment of the presentinvention. As explained below, controller 300 can be implemented inhardware, software, and/or firmware, depending upon the design choicesof the system designer.

[0041] Potential applications for an embodiment of the present inventioninclude, for example, employment in a boost starter converter for a fuelcell system and potential use in HEVs. Another potential application foran embodiment of the present invention is employment for abi-directional converter for similar applications. Simply replacingdiodes D₆ 34 and D₈ 38 as shown in FIG. 3 with transistors S₆ 100 and S₈102 achieves bi-directional operation, as illustrated in FIG. 5,according to an embodiment of the present invention. Controller unit 500controls switches S₁-S₈ in the manner described herein, according to anembodiment of the present invention. Controller unit 500 interacts withmemory unit 502, in one embodiment, to retrieve instruction code whicheffects part or all of the control processes described herein.

[0042] Those having ordinary skill in the art will recognize that thestate of the art has progressed to the point where there is littledistinction left between hardware and software implementations ofaspects of systems; the use of hardware or software is generally (butnot always, in that in certain contexts the choice between hardware andsoftware can become significant) a design choice representing cost vs.efficiency tradeoffs. Those having ordinary skill in the art willappreciate that there are various vehicles by which aspects of processesand/or systems described herein can be effected (e.g., hardware,software, and/or firmware), and that the preferred vehicle will varywith the context in which the processes and/or systems are deployed. Forexample, if an implementer determines that speed and accuracy areparamount, the implementer may opt for a hardware and/or firmwarevehicle; alternatively, if flexibility is paramount, the implementer mayopt for a solely software implementation; or, yet again alternatively,the implementer may opt for some combination of hardware, software,and/or firmware. Hence, there are several possible vehicles by whichaspects of the processes described herein may be effected, none of whichis inherently superior to the other in that any vehicle to be utilizedis a choice dependent upon the context in which the vehicle will bedeployed and the specific concerns (e.g., speed, flexibility, orpredictability) of the implementer, any of which may vary.

[0043] The foregoing detailed description has set forth variousembodiments of the devices and/or processes via the use of blockdiagrams, schematics, and examples. Insofar as such block diagrams,schematics and examples contain one or more functions and/or operations,it will be understood by those within the art that each function and/oroperation within such block diagrams, flowcharts, or examples can beimplemented, individually and/or collectively, by a wide range ofhardware, software, firmware, or virtually any combination thereof. Inone embodiment, the present invention may be implemented via ApplicationSpecific Integrated Circuits (ASICs). However, those skilled in the artwill recognize that the embodiments disclosed herein, in whole or inpart, can be equivalently implemented in standard Integrated Circuits,as one or more computer programs running on one or more computers (e.g.,as one or more programs running on one or more computer systems), as oneor more programs running on one or more controllers (e.g.,microcontrollers) as one or more programs running on one or moreprocessors (e.g., microprocessors), as firmware, or as virtually anycombination thereof, and that designing the circuitry and/or writing thecode for the software and or firmware would be well within the skill ofone of ordinary skill in the art in light of this disclosure. Inaddition, those skilled in the art will appreciate that the mechanismsof the present invention are capable of being distributed as a programproduct in a variety of forms, and that an illustrative embodiment ofthe present invention applies equally regardless of the particular typeof signal bearing media used to actually carry out the distribution.Examples of signal bearing media include, but are not limited to, thefollowing: recordable type media such as floppy disks, hard disk drives,CD ROMs, digital tape, and computer memory; and transmission type mediasuch as digital and analogue communication links using TDM or IP basedcommunication links (e.g., packet links).

[0044] Various preferred embodiments of the invention have beendescribed in fulfillment of the various objects of the invention. Itshould be recognized that these embodiments are merely illustrative ofthe principles of the present invention. Numerous modifications andadaptations thereof will be readily apparent to those skilled in the artwithout departing from the spirit and scope of the present invention.

1. An isolated boost DC/DC converter, comprising: an inductor capable ofstoring energy from a direct current power source; an inverter circuitcoupled to the inductor and capable of converting direct current toalternating current; a transformer coupled to the inverter circuit, thetransformer comprising a primary and a secondary and capable of steppingup voltage of the alternating current; a rectifier circuit coupled tothe transformer capable of converting the stepped up alternating currentto direct current for delivery to a load; and a programmable controllerprogrammed with a commutation logic that, in a charging inductorswitching mode of the inverter circuit between alternating positive andnegative current switching modes of the inverter circuit, causes energyfrom the direct current power source to be stored in the inductor, andcauses a leakage inductance current of the transformer to be preset to avalue that approaches a value of an input inductor current by shortcircuiting the secondary during the charging inductor switching mode ofthe inverter circuit.
 2. The isolated boost DC/DC converter according toclaim 1 wherein the inverter circuit further comprises first, second,third and fourth switches, and the programmable controller is programmedwith the commutation logic that, in the charging inductor switching modeof the inverter circuit, causes the primary to be short circuited byturning on the first, second, third and fourth switches.
 3. The isolatedboost DC/DC converter according to claim 2 wherein the programmablecontroller is programmed with the commutation logic that, in thepositive current switching mode, turns on the first and second switchesand turns off the third and fourth switches, such that energy istransferred from the inductor to the secondary, and current in thetransformer is positive.
 4. The isolated boost DC/DC converter accordingto claim 3 wherein the programmable controller is programmed with thecommutation logic that, in the negative current switching mode, turns onthe third and fourth switches and turns off the first and secondswitches, such that energy is transferred from the inductor to thesecondary, and current in the transformer is negative.
 5. The isolatedboost DC/DC converter according to claim 1 wherein the rectifier circuitfurther comprises fifth, sixth, seventh, and eighth switches, andwherein the programmable controller is programmed with the commutationlogic that causes the leakage inductance current of the transformer tobe preset to the value that approaches the value of the input inductorcurrent and causes commutation of the leakage inductance to beaccelerated by turning on one of the fifth and seventh switches to shortcircuit the secondary during the charging inductor switching mode of theinverter circuit.
 6. The isolated boost DC/DC converter according toclaim 5 wherein the sixth and eighth switches further comprise diodes,and wherein the eighth switch is automatically on when the fifth switchis turned on, and the sixth switch is automatically on when the seventhswitch is turned on.
 7. The isolated boost DC/DC converter according toclaim 6 wherein the programmable controller is programmed with thecommutation logic that causes the secondary to be short circuited byturning on one of the fifth switch and the seventh switch of therectifier circuit during the positive and negative current switchingmodes to create a momentary short circuit on the secondary of thetransformer to preset the leakage inductance current of the transformerto a predetermined value that approaches the current value in theinductor.
 8. The isolated boost DC/DC converter according to claim 5wherein the programmable controller is programmed with the commutationlogic that turns on the fifth and seventh switches according to apredetermined time sequence to create a zero voltage soft switchingcondition.
 9. The isolated boost DC/DC converter according to claim 8wherein the programmable controller is programmed with the commutationlogic that turns on the fifth and seventh switches into a conductingdiode to create the zero voltage soft switching condition.
 10. Theisolated boost DC/DC converter according to claim 9 wherein theprogrammable controller is programmed with the commutation logic thatcauses the fifth and seventh switches to each be turned on into theconducting diode.
 11. The isolated boost DC/DC converter according toclaim 5 wherein the programmable controller is programmed with thecommutation logic that causes one of the fifth and seventh switches tobe turned on before the charging inductor switching mode of the invertercircuit.
 12. The isolated boost DC/DC converter according to claim 1,wherein the rectifier circuit further comprises fifth, sixth, seventh,and eighth switches, wherein the sixth and seventh switches furthercomprise diodes, wherein the seventh switch is automatically on when thefifth switch is turned on, and the sixth switch is automatically on whenthe eighth switch is turned on, and wherein the programmable controlleris programmed with the commutation logic that causes the leakageinductance current of the transformer to be preset to the value thatapproaches the value of the input inductor current and commutation ofthe leakage inductance current is accelerated by turning on one of thefifth and eighth switches to short circuit the secondary during thecharging inductor switching mode of the inverter circuit.
 13. Theisolated boost DC/DC converter according to claim 1 wherein therectifier circuit further comprises fifth, sixth, seventh, and eighthswitches, wherein the fifth and eighth switches further comprise diodes,wherein the eighth switch is automatically on when the sixth switch isturned on, and the fifth switch is automatically on when the seventhswitch is turned on, and wherein the programmable controller isprogrammed with the commutation logic that causes the leakage inductancecurrent of the transformer to be preset to the value that approaches thevalue of the input inductor current and commutation of the leakageinductance current is accelerated by turning on one of the sixth andseventh switches to short circuit the secondary during the charginginductor switching mode of the inverter circuit.
 14. The isolated boostDC/DC converter according to claim 1 wherein the rectifier circuitfurther comprises fifth, sixth, seventh, and eighth switches, whereinthe fifth and seventh switches further comprise diodes, wherein theseventh switch is automatically on when the sixth switch is turned on,and the sixth switch is automatically on when the eighth switch isturned on, and wherein the programmable controller is programmed withthe commutation logic that causes the leakage inductance current of thetransformer to be preset to the value that approaches the value of theinput inductor current and commutation of the leakage inductance currentis accelerated by turning on one of the sixth and eighth switches toshort circuit the secondary during the charging inductor switching modeof the inverter circuit.
 15. The isolated boost DC/DC converteraccording to claim 1 wherein the rectifier circuit further comprisesfifth, sixth, seventh, and eighth switches, each of which is atransistor capable of bi-directional operation.
 16. A method forcontrolling commutation of an isolated boost DC/DC converter,comprising: storing energy from a direct current power source by aninductor during a charging inductor mode; converting direct current toalternating current by an inverter circuit coupled to the inductor,stepping up the voltage of the alternating current by a transformercoupled to the inverter circuit comprising a primary and a secondary;and converting the stepped up alternating current to direct current by arectifier circuit coupled to the transformer for delivery to a load;wherein the energy from the direct current power source is stored in theinductor in a charging inductor switching mode of the inverter circuitbetween alternating positive and negative current switching modes of theinverter circuit, and a leakage inductance current of the transformer ispreset to a value approaches a value of an input inductor current byshort circuiting the secondary during the charging inductor switchingmode of the inverter circuit, according to a pre-programmed commutationlogic of a programmable controller.
 17. A controller programmed tocontrol communication of an isolated boost DC/DC converter comprising atransformer with a primary and secondary, by: in a charging inductorswitching mode of an inverter circuit between alternating positive andnegative current switching modes of the inverter circuit, short circuitsthe secondary to cause energy from a direct current power source to bestored in an inductor and to preset a leakage inductance current of thetransformer to a value that approaches a value of an input inductor. 18.The controller of claim 17 further comprising four switches forming abridge coupled to the secondary, and wherein each of the four switchesare in a conducting state during a same interval to short circuit thesecondary.
 19. A device for use with a DC/DC converter, the devicecomprising: a transformer comprising a primary winding and a secondarywinding; and a switch controlled to short circuit the secondary windingduring a reversal of current direction through the primary winding. 20.The device of claim 19 wherein the switch comprises a transistorcomprising an associated capacitance.
 21. The device of claim 19 whereinthe switch controlled to short circuit the secondary winding during areversal of current direction through the primary winding furthercomprises: a first switch controlled to couple a first end of thesecondary winding to a first node substantially subsequent to a firsttime when a positive polarity voltage is applied between a first end ofthe primary winding and a second end of the primary winding; a firstdiode comprising a first-diode anode (+) coupled to a second end of thesecondary winding and a first-diode cathode (−) coupled to the firstnode; and a second diode comprising a second-diode cathode (−) coupledto the second end of the secondary winding and a first-diode anode (+)coupled to a second node.
 22. The device of claim 21, furthercomprising: at least one of the first diode and the second diodecomprising an associated capacitance.
 23. The device of claim 21,further comprising: a second switch controlled to couple a terminal ofan inductor to the first end of the primary winding during the firsttime; and a third switch controlled to couple the second end of theprimary winding to a third node during the first time.
 24. The device ofclaim 23, further comprising: a fourth switch controlled to couple thefirst end of the secondary winding to the second node substantiallysubsequent to a second time when a negative polarity voltage is appliedbetween the first end of the primary winding and the second end of theprimary winding; a fifth switch controlled to couple the terminal of theinductor to the second end of the primary winding during the secondtime; and a sixth switch controlled to couple the first end of theprimary winding to the third node during the second time.
 25. A methodfor use with a DC/DC converter, the method comprising: applying a firstpolarity voltage across a first end and a second end of a primarywinding of a transformer during a time interval; and shorting asecondary winding of the transformer during at least one of a timesubstantially contemporaneous with the end of the time interval and atime substantially subsequent to the end of the time interval.
 26. Themethod of claim 25 wherein the shorting a secondary winding of thetransformer comprises maintaining a connection between a first end ofthe secondary winding and a first node subsequent to an application ofthe first polarity voltage to the primary winding.
 27. The method ofclaim 26 wherein the maintaining a connection between a first end of thesecondary winding and a first node subsequent to an application of thefirst polarity voltage to the primary winding comprises maintaining theconnection for a time sufficient such that a current in the secondarywinding reverses direction relative to a direction of current in thesecondary winding during the time interval.
 28. A method for use with aDC/DC converter, the method comprising: closing a first switch couplinga first end of a secondary transformer with a first node at a first timeprior to a second time at which application of a first polarity voltageto a primary transformer causes a forward biasing voltage of a firstdiode comprising a first-diode cathode (−) coupled to a second end ofthe secondary transformer and a first-diode anode (+) coupled to asecond node to be substantially exceeded.
 29. The method of claim 28,further comprising closing a second switch coupling the first end of thesecondary transformer with the second node at a third time prior to afourth time at which application of a second polarity voltage to theprimary transformer causes a forward biasing voltage of a second diodecomprising a second-diode anode (+) coupled to the second end of thesecondary transformer and a second-diode cathode (−) coupled to thefirst node to be substantially exceeded.